Programmable transmit continuous-time filter

ABSTRACT

A programmable-current transmit continuous-time filter (TX-CTF) system can be included in a radio frequency (RF) transmitter. The input of the TX-CTF can receive a baseband transmission signal, and the output of the TX-CTF can be provided to an upconversion mixer for conversion to RF for transmission. The TX-CTF includes amplifier circuitry and passive circuitry that together define the filter parameters. The TX-CTF further includes programmable current circuitry that provides a programmable bias current to the amplifier circuitry. The TX-CTF system also includes control logic that receives one or more transmitter control signals and, in response, generates signals that control the bias current provided to the TX-CTF.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/730,108, filed Jun. 3, 2015, which is a continuation of U.S. patentapplication Ser. No. 14/183,805, filed Feb. 19, 2014, which is acontinuation of U.S. patent application Ser. No. 13/185,391, filed Jul.18, 2011, which is a continuation of International Application No.PCT/US2009/054417, filed Aug. 20, 2009, which claims the benefit of thefiling date of U.S. Provisional Patent Application No. 61/145,638, filedJan. 19, 2009, the benefits of the filing dates of which are herebyclaimed and the specifications of which are incorporated herein by thisreference.

BACKGROUND

A transmit continuous-time filter (TX-CTF) is a frequency-selectivecircuit that is typically included in the transmitter portion of sometypes of cellular telephones (also referred to as handsets). A TX-CTFtypically receives the output of a digital-to-analog converter (DAC) andattenuates the DAC aliasing and noise. The output of the TX-CTF istypically provided to an active upconversion mixer that upconverts thesignal from a baseband frequency to the desired radio frequency (RF)band for transmission.

Some tri-mode cellular handsets support Wideband Code Division MultipleAccess (WCDMA) modulation, the Gaussian Minimum Shift Keying (GMSK)modulation used in the Global System for Mobile telecommunication (GSM)standard, and the 8-Phase Shift Keying (8PSK) modulation used in theEnhanced Data Rates for Global Evolution (EDGE) standard (also known asthe Enhanced Data Rates for GSM Evolution standard). Enabling all threeof the above modes in the same handset imposes stringent performancerequirements on the TX-CTF, including high current drive capability,high linearity, low input referred noise, and low bandpass ripple.Designing a TX-CTF that meets all of these requirements can beproblematic. Straightforward solutions to these challenges that may beproposed, such as increasing current, can introduce other problems. Forexample, high TX-CTF current consumption can lead to an undesirably highcurrent drain on the cellular handset battery. Minimizing batterycurrent drain is desirable so that talk time and standby time (i.e., theamount of time the handset can be used before the battery requiresrecharging) can be maximized and battery size can be minimized.

A typical TX-CTF 10 is shown in FIG. 1. As a typical cellular handsettransmitter uses a form of quadrature modulation, TX-CTF 10 includes anin-phase (I) portion 12 and a quadrature (Q) portion 14. As portions 12and 14 are essentially identical, only portion 12 is described in detailherein. Portion 12 includes two sections 16 and 18. Section 16 can be,for example, a 2^(nd)-order biquadratic stage, and section 18 can be,for example, a 1^(st)-order real pole stage. Section 16 includes a firstamplifier 20 as well as passive circuitry that can include, for example,capacitors 22, 24 and 26, and resistors 28, 30, 32, 34, 36 and 38. Thepassive circuitry can be selected and connected to first amplifier 20 inan arrangement that defines the desired filter parameters, such as thefilter poles and/or zeroes that characterize a 2^(nd)-order biquadraticfilter. Similarly, section 18 includes a second amplifier 40 as well aspassive circuitry that can include, for example, capacitors 42 and 44,and resistors 46, 48, 50 and 52. The passive circuitry of section 18likewise can be selected and connected to second amplifier 40 in anarrangement that defines the desired filter parameters, such as thefilter poles and/or zeroes that characterize a 1^(st)-order real polefilter.

In operation, a differential input signal V1 (the negative side of whichis represented in FIG. 1 as “V1_N,” and the positive side as “V1_P”) isprovided to stage 16, which outputs a signal V2 (the negative side ofwhich is represented as “V2_N,” and the positive side as “V2_P”). Thesignal V2 is in turn provided to stage 18, which outputs a signal V3(the negative side of which is represented as “V3_N” and the positive as“V3_P”).

As in a typical cellular handset, the in-phase (I) and quadrature (Q)outputs of TX-CTF 10 are provided to active upconversion mixers 54 and56, respectively. Each of upconversion mixers 54 and 56 is typically ofthe Gilbert cell type, which presents a high-impedance load to TX-CTF10. The TX-CTF 10 is readily capable of driving the high impedance loadwith low current and maintaining the required linearity.

It would be desirable to provide a TX-CTF for a multi-mode cellularhandset that can meet the above-described performance criteria orsimilar performance criteria without consuming excessive current.

SUMMARY

Embodiments of the invention relate to a programmable-current transmitcontinuous-time filter (TX-CTF) system in a radio frequency (RF)transmitter, and a method of transmitter operation. The input of theTX-GTE can receive a baseband transmission signal, and the output of theTX-GTE can be provided to an upconversion mixer for conversion to RF fortransmission. The TX-CTF includes amplifier circuitry and passivecircuitry that together define the filter parameters. The TX-CTF furtherincludes programmable current circuitry that provides a programmablebias current to the amplifier circuitry. The TX-CTF system also includescontrol logic that receives one or more transmitter control signals and,in response, generates signals that control the bias current provided tothe TX-CTF. The transmitter control signals can include, for example,one or more of the following: a transmitter modulation mode signal, atransmitter band signal, and a transmitter power signal.

Other systems, methods, features, and advantages of the invention willbe or become apparent to one with skill in the art upon examination ofthe following figures and detailed description.

BRIEF DESCRIPTION OF THE FIGURES

The invention can be better understood with reference to the followingfigures. The components within the figures are not necessarily to scale,emphasis instead being placed upon clearly illustrating the principlesof the invention. Moreover, in the figures, like reference numeralsdesignate corresponding parts throughout the different views.

FIG. 1 is a block diagram showing a known or prior transmitcontinuous-time filter (TX-CTF) connected to an active upconversionmixer.

FIG. 2 is a block diagram of an exemplary cellular handset that includesa transmitter portion having a TX-CTF system in accordance an exemplaryembodiment of the invention.

FIG. 3 is a block diagram of the exemplary transmitter portion of FIG.2.

FIG. 4 is a block diagram of the exemplary TX-CTF system of FIG. 3.

FIG. 5 is a block diagram of the exemplary second amplifier system ofthe TX-CTF system of FIG. 4.

FIG. 6A is a flow diagram illustrating exemplary decision logic of thesecond amplifier system of FIG. 5.

FIG. 6B is a continuation of the flow diagram of FIG. 6A.

FIG. 7 is a block diagram of a bias control logic portion of theexemplary first stage bias generator circuitry of the second amplifiersystem of FIG. 5.

FIG. 8 is a block diagram of an PFET signal generator portion of theexemplary first stage bias generator circuitry of the second amplifiersystem of FIG. 5.

FIG. 9 is a block diagram of a NFET signal generator portion of theexemplary first stage bias generator circuitry of the second amplifiersystem of FIG. 5.

FIG. 10 is a block diagram of a programmable-current first stage of thesecond amplifier system of FIG. 5.

FIG. 11 is a block diagram of a programmable-current second stage of thesecond amplifier system of FIG. 5.

FIG. 12 is a table showing an exemplary relation between a controlsignal and resulting first stage bias current.

FIG. 13 is a table showing an exemplary relation between another controlsignal and resulting second stage bias current.

DETAILED DESCRIPTION

As illustrated in FIGS. 2-3, in an illustrative or exemplary embodimentof the invention, a mobile wireless telecommunication device 58, such asa cellular telephone handset, includes a radio frequency (RF) subsystem60, an antenna 62, a baseband subsystem 64, and a user interface section66. The RF subsystem 60 includes a transmitter portion 68 and a receiverportion 70. The output of transmitter portion 68 and the input ofreceiver portion 70 are coupled to antenna 62 via a front-end module 72that allows simultaneous passage of both the transmitted RF signalproduced by transmitter portion 68 and the received RF signal that isprovided to receiver portion 70. But for certain elements of transmitterportion 68 described below, the above-listed elements can be of thetypes conventionally included in such mobile wireless telecommunicationdevices. As conventional elements, they are well understood by personsof ordinary skill in the art to which the present invention relates and,accordingly, not described in further detail herein. However, unlikeconventional transmitter portions of such mobile wirelesstelecommunication devices, transmitter portion 68 includes a transmitcontinuous-time filter (TX-CTF) system 74 (FIG. 3), having the featuresdescribed below.

Transmitter portion 68 receives as an input a digital baseband signalfrom baseband subsystem 64 (FIG. 1) and outputs an RF signal to betransmitted. Transmitter portion 68 further includes a digital-to-analogconverter 76, a dual-mode modulator and upconversion mixer 78, and apower amplifier system 80. Digital-to-analog converter 76 converts thedigital baseband signal to analog form and provides the resulting analogbaseband signal to TX-CTF system 74. The output of TX-CTF system 74 isprovided to dual-mode modulator and upconversion mixer 78. The output ofdual-mode modulator and upconversion mixer 78 is provided to poweramplifier system 80.

Baseband subsystem 64, through an internal microprocessor system orsimilar logic (not separately shown), can control various operationalaspects of mobile wireless telecommunication device 58. For example,baseband subsystem 64 can produce one or more transmitter controlsignals on one or more connections 82 (e.g., a digital bus) that affectthe operation of transmitter portion 68. Transmitter control signals caninclude, for example, one or more of the following: a transmittermodulation mode signal, a transmitter band signal, and a transmitterpower signal.

In the exemplary embodiment, a transmitter modulation mode signal caninstruct modulator and upconversion mixer 78 to operate in a selectedone of at least two modulation modes. As well understood in the art,multi-mode (e.g., dual-mode or tri-mode) cellular handsets enableroaming between geographic regions in which cellular telecommunicationstandards differ. Although in other embodiments there can be more thantwo modes, in this exemplary embodiment the modulation modes can includethe WCDMA modulation that is associated with the WCDMA standard and theGMSK modulation that is associated with GSM standard and some aspects ofthe EDGE standard. As well understood in the art, the EDGE standard usesGMSK modulation for the lower four of its nine modulation and codingschemes but uses higher-order 8PSK modulation for the upper five ofthose nine modulation and coding schemes.

In a manner well understood in the art, in response to a transmittermodulation mode signal representing a command or instruction issued bybaseband subsystem 64 to operate in WCDMA mode, transmitter portion 68modulates the signal to be transmitted in accordance with the WCDMAstandard. In response to a transmitter modulation mode signalrepresenting a command or instruction issued by baseband subsystem 64 tooperate in EDGE modulation mode, transmitter portion 68 modulates thesignal to be transmitted in accordance with the EDGE standard, i.e.,either GMSK or 8PSK modulation. In response to a transmitter modulationmode command issued by baseband subsystem 64 to operate in GSM mode,transmitter portion 68 modulates the signal to be transmitted inaccordance with the GSM standard, i.e., GMSK modulation. Although in theexemplary embodiment the modes include any two or more of GSM, EDGE andWCDMA, in other embodiments the modes selected in response to thetransmitter modulation mode signal can include these or any othermodulation modes known in the art.

The transmitter band signal can instruct modulator and upconversionmixer 78 to operate in a selected one of two or more (frequency) bands.Although in other embodiments there can be more than two bands, in thisexemplary embodiment there is a high band and a low band. As wellunderstood in the art, dual-band, tri-band, quad-band, etc., cellularhandsets enable operation in geographic regions in which cellulartelecommunication standards specify different frequency bands. Inresponse to a transmitter band signal representing a command issued bybaseband subsystem 64 to operate in a selected frequency band,transmitter portion 68 upconverts the signal to be transmitted to theselected frequency band.

The transmitter power signal can include one or more signals thatindicate the output power at which transmitter portion 68 is operating,indicate the output power at which transmitter portion 68 is instructedto operate, or relate in any other way to the transmitted RF signalpower. For example, baseband subsystem 64 can issue a power controlcommand that instructs power amplifier system 80 to set its gain to aselected value and thus amplify its input signal to a correspondingpower level. Control circuitry 84 in power amplifier system 80 providesa closed-loop feedback system that maintains the power amplifier 86 atthe selected power level or makes other adjustments in response to otherconditions, as well understood in the art. There can be any number ofselectable power levels, but for illustrative purposes herein there canbe two selectable power levels: a low power level and a high powerlevel. As persons of skill in the art understand, the values of the lowand high power levels can be in accordance with the applicable standards(e.g., WCDMA, EDGE, GSM, etc.), but for illustrative purposes herein itneed only be understood that there are at least two different selectablepower levels.

Another transmitter power signal that can be included in addition oralternatively to the transmitter power control signal described abovecan indicate a measured or detected power level at which transmitterportion 68 is actually operating. Signals 88 indicating such a measuredpower level at which transmitter portion 68 is operating are produced incontrol circuitry 84 as part of the closed-loop feedback power controlprocess.

Another transmitter power signal that can be included in addition oralternatively to the transmitter power control signal described abovecan indicate a “back-off condition in which transmitter operationtransitions from a higher transmit power to a lower transmit power.Although not shown for purposes of clarity, transmitter portion 68 canback its transmit power off by, for example, switching an attenuatingcircuit into the signal path. This type of back-off scheme iscontemplated by the WCDMA standard. However, in embodiments of theinvention the transmitted power control signal can indicate such a WCDMAback-off condition or any other similar type of back-off condition.

The TX-CTF 74 system receives one or more of the above-describedtransmitter control signals and, in response, adjusts the bias currentprovided to its amplifier circuitry, as described in further detailbelow.

As illustrated in. FIG. 4, in the exemplary embodiment TX-CTF system 74includes an in-phase (I) portion 92 and a quadrature (Q) portion 94. Asin-phase portion 92 and quadrature 94 are essentially identical, onlyin-phase portion 92 is described in detail herein. In-phase portion 92includes first and second sections 96 and 98. Section 96 can define, forexample, a 2^(nd)-order biquadratic filter, and section 98 can define,for example, a 1^(st)-order real pole filter. Section 96 includes afirst amplifier 100 as well as passive circuitry that can include, forexample, capacitors 102, 104 and 106, and resistors 108, 110, 112, 114,116 and 118. The passive circuitry can be selected and connected tofirst amplifier 100 in an arrangement that defines the desired filterparameters, such as the filter poles and zeroes that characterize a2^(nd)-order biquadratic filter. Similarly, section 98 includes a secondamplifier system 120 as well as passive circuitry that can include, forexample, capacitors 122 and 124, and resistors 126, 128, 130 and 132.The passive circuitry of section 98 can be selected and connected tosecond amplifier system 120 in an arrangement that defines the desiredfilter parameters, such as the filter poles and zeroes that characterizea 1^(st)-order real pole filter. Although in the exemplary embodimentsection 96 defines a 2^(nd)-order biquadratic filter and section 98defines a 1^(st)-order real pole filter, in other embodiments eithersection can comprise any other suitable type of continuous-time filter.

In operation, a differential input signal V1 (the negative side of whichis represented in FIG. 4 as “V1_N,” and the positive polarity as “V1_P”)is provided to section 96, which outputs a signal V2 (the negativepolarity of which is represented as “V2_N,” and the positive polarity as“V2_P”). The signal V2 is in turn provided to section 98, which outputsa signal V3 (the negative polarity of which is represented as “V3_N,”and the positive polarity as “V3_P”).

The in-phase (I) and quadrature (Q) outputs of TX-CTF system 74 areprovided to modulator and upconversion mixer 78 (FIG. 3). In theexemplary embodiment, modulator and upconversion mixer 78 includespassive, rather than active, upconversion mixer circuitry. Passivemixers consume less current and can operate at lower supply voltagesthan active mixers. Using passive mixers is therefore beneficial inreducing power consumption and also in implementing the circuit in lowersupply voltages, which permits a smaller handset battery. Like activeupconversion mixer circuitry, the passive upconversion mixer circuitryof modulator and upconversion mixer 78 performs frequency translation ofthe signal, centering the signal at an RF carrier frequency. However, inpassive upconversion mixer circuitry, the frequency translation, i.e.,upconversion, also leads to a potentially disadvantageous reduction inthe input impedance of modulator and upconversion mixer 78. Therefore,for TX-CTF system 74 to drive modulator and upconversion mixer 78 withadequate power levels, TX-CTF system 74 is provided with substantiallylower output impedance than the prior TX-CTF 10 (FIG. 1). In the mannerdescribed below, TX-CTF system 74 provides such lower output impedancewithout sacrificing the linearity required by the GMSK modulation andwithout consuming excessive current.

It can also be noted that use of passive upconversion mixer circuitryleads to substantially more noise being transferred from TX-CTF system74 to the RF output than would be transferred using active upconversionmixer circuitry. Such noise emanating from transmitter portion 68 (FIG.2) can undesirably couple into receiver portion 70. A high degree oftransmitter noise can impede proper transceiver operation and can exceedpermissible thresholds established by the WCDMA, GSM, EDGE or otherstandards. However, in some instances of handset operation, such asoperation in WCDMA mode, the permissible threshold of noise establishedby the WCDMA standard depends on the transmit power. For example, whenthe handset is transmitting at a high power level, a lower threshold ofnoise is permitted. Conversely, when the handset is transmitting at alow power level, a higher threshold of noise is permitted. In addition,when the transmitter is operating in the above-described WCDMA back-ofcondition, i.e., the transmitter power is being reduced from a higherlevel to a lower level, the permitted noise threshold is raised as thepower is lowered. One example of such a rise in permitted noisethreshold versus transmitter power is that, for every 1 decibel of powerreduction, the permitted noise threshold is raised by one decibel. Lownoise in TX-CTF system 74 requires high current. Conversely, high noisein TX-CTF system 74 can be obtained with lower current. In the mannerdescribed below, TX-CTF system 74 promotes minimization of currentconsumption by providing the lower noise thresholds in response tohandset operating conditions, such as the low, high, and back-offtransmitter power conditions described above.

Second amplifier system 120 (FIG. 4) of in-phase (I) portion 92 isillustrated in further detail in FIG. 5. Although quadrature (Q) portion94 includes another such second amplifier system, this other secondamplifier system is identical to second amplifier system 120 and istherefore not described herein.

As illustrated in FIG. 5, second amplifier system 120 includes aprogrammable-current first stage 134, a programmable-current secondstage 136, and decision logic 138. In the illustrated embodiment,decision logic 138 receives the above-described transmitter controlsignals and, in response, controls or programs the bias current providedto amplifier circuitry (described below) of programmable-current firststage 134 and programmable-current second stage 136. Decision logic 138can include any suitable logic that enables a determination of theamount of bias current to provide. For example, decision logic 138 caninclude a microprocessor or digital signal processor (not shown)programmed in accordance with the logic represented by the flow diagramof FIGS. 6A-B. One output of decision logic 138, representing the amountof bias current to provide to programmable-current first stage 134, isrepresented in the exemplary embodiment by a 3-bit digital word orsignal, ISET, comprising bits ISET0, ISET1 and ISET2. Another output ofdecision logic 138, representing the amount of bias current to provideto programmable-current second stage 136, is represented in theexemplary embodiment by a 3-bit digital word or signal, AB, comprisingbits AB0, AB1 and AB2. (The signal name “AB” is a reference to the ABclass of the amplifier circuitry in programmable-current second stage136 in the exemplary embodiment.) Inverter logic 139 provides the signalAB along with its complement to programmable-current second stage 136.

As illustrated in FIGS. 6A-B, decision logic 138 can, for example,include several logical sections relating to the operation ofprogrammable-current first stage 134 and programmable-current secondstage 136 in response to handset operating conditions. It should berecalled from the description above with regard to FIG. 4 thatprogrammable-current first stage 134 and programmable-current secondstage 136 are included in second amplifier system 120 of second section98. As described in further detail below with regard to FIGS. 6A-B, afirst logical section 139 determines the amount of bias current toprovide to programmable-current first stage 134 in response tocombinations of transmitter modulation mode and transmitter power; asecond logical section 141 determines the amount of bias current toprovide to programmable-current second stage 136 in response totransmitter band; a third logical section 143 determines the amount ofbias current to provide to one or both of programmable-current firststage 134 and programmable-current second stage 136 in response tocombinations of transmitter power and whether a transmitter WCDMA powerback-off condition exists; and a fourth logical section 145 determinesthe amount of bias current to provide to programmable-current secondstage 136 in response to transmitter modulation mode. Although theselogical sections are shown and described for purposes of clarity in theexemplary embodiment as being sequential, they can be integrated withone another in any other suitable manner. For example, in otherembodiments such logical sections can operate in parallel or,alternatively, can be combined in the form of a single logicaloperation, such as evaluation of a formula in software or a network oflogic in hardware.

The contributions indicated by logical sections 139, 141, 143 and 145 tothe total amount of bias current to be provided to programmable-currentfirst stage 134 and the total amount of bias current to be provided toprogrammable-current second stage 136 can be added together to determinethe totals. The logic represented by the flow diagram of FIGS. 6A-B isintended to be merely exemplary, and other suitable logic will occurreadily to persons skilled in the art to which the invention relates inview of the teachings herein. It should also be noted that the logicrepresented by the flow diagram of FIGS. 6A-B is shown in isolation fromother transceiver operations for purposes of clarity. Persons skilled inthe art understand that such logic is to be applied at appropriatetimes, such as upon the occurrence of changes in operating conditions inthe transceiver.

With regard to logical section. 139, if the transmitter modulation modeis WCDMA, as indicated by block 140, and if a transmitter power signalindicates that the transmitter power is low, as indicated by block 142,then decision logic 138 outputs the ISET signal to adjust the biascurrent provided to programmable-current first stage 134 to a lower ordecreased level (with respect to some predetermined range or scale inwhich bias current levels can be programmed), as indicated by block 144.For example, as described above, the bias current can be adjusted to alower or decreased level by contributing a smaller number (selected froma predetermined range or scale of numbers) to the digital sum that theISET signal indicates. If the transmitter modulation mode is WCDMA, asindicated by block 140, and if a transmitter power signal indicates thatthe transmitter power is high, as indicated by block 142, then decisionlogic 138 outputs the ISET signal to adjust the bias current provided toprogrammable-current first stage 134 to a higher or increased level, asindicated by block 146. For example, as described above, the biascurrent can be adjusted to a lower level by contributing a larger numberto the digital sum that the ISET signal represents.

With regard to logical section 141 section, if the transmitter isoperating in the low frequency band, as indicated by block 148, thendecision logic 138 outputs the AB signal to adjust the bias currentprovided to programmable-current second stage 136 to a lower ordecreased level, as indicated by block 150. If the transmitter is notoperating in the low frequency band (i.e., it is operating in the highfrequency band), then decision logic 138 outputs the AB signal to adjustthe bias current provided to programmable-current second stage 136 to ahigher or increased level, as indicated by block 152.

With regard to logical section 143, if the transmitter is operating in apower back-off condition, as indicated by block 154, then decision logic138 outputs a combination of one or both of the ISET signal and the ABsignal to adjust the bias current provided to one or both ofprogrammable-current first stage 134 and programmable-current secondstage 136, respectively, to a lower or decreased level, as indicated byblock 156.

Furthermore, in some embodiments, after the bias current has beenadjusted in this manner but while the transmitter remains in the powerback-off condition, decision logic 138 can further change the ISETsignal to adjust the bias current provided to one or both ofprogrammable-current first stage 134 and programmable-current secondstage 136 to a different bias current level, i.e., a differentcombination of bias current levels provided to programmable-currentfirst stage 134 and programmable-current second stage 136. For example,decision logic 138 could initially cause only the bias current providedto programmable-current first stage 134 to be decreased and not causethe bias current provided to programmable-current second stage 136 to bedecreased. At some point in time thereafter (e.g., on the order ofmilliseconds), while the transmitter remains in the power back-offcondition, decision logic 138 could then cause both the bias currentprovided to programmable-current first stage 134 and the bias currentprovided to programmable-current second stage 136 to be furtherdecreased or otherwise adjusted. Then, at a still later point in timebut still while the transmitter remains in the power backoff condition,decision logic 138 could further decrease or otherwise adjust only thebias current provided to programmable-current second stage 136 and notfurther decrease or otherwise adjust the bias current provided toprogrammable-current first stage 134. The foregoing adjustment sequenceis intended only as an example, and others will occur to persons skilledin the art in view of the teachings herein.

With regard to logical section 145, if the transmitter is operating inthe WCDMA modulation mode, as indicated by block 162, then decisionlogic 138 outputs the AB signal to adjust the bias current provided toprogrammable-current second stage 136 to a lower or decreased level, asindicated by block 164. If the transmitter is not operating in the WCDMAmodulation mode, then decision logic 138 outputs the AB signal to adjustthe bias current provided to programmable-current second stage 136 to ahigher or increased level, as indicated by block 166.

In response to the ISET signal, bias control logic 168 (FIG. 5) insecond amplifier system 120 produces a PFET (p-channel field-effecttransistor) digital control word or signal, PCTRL, comprising bitsPCTRL2, PCTRL3, PCTRL4, PCTRL5, PCTRL6 and PCTRL7, as well as an NFET(n-channel field-effect transistor) digital control word or signal,NCTRL, comprising bits NCTRL2, NCTRL3, NCTRL4, NCTRL5, NCTRL6 andNCTRL7. As the portion of bias control logic 168 that produces thesignal NCTRL is identical to the portion that produces the signal PCTRL,only the portion that produces the signal NCTRL is illustrated anddescribed herein. It should be noted that the number of bits in thevarious signals described herein are merely exemplary, and that in otherembodiments such signals can have any other suitable number of bits.

As illustrated in FIG. 7, the portion of bias control logic 168 thatproduces the signal NCTRL can include, for example, a network ofcombinational logic, such as: inverters 170, 172, 174, 176, 178, 180,182, 184, 186, 188 and 190; NOR gates 192, 194 and 196; a NAND gate 198;an OR gate 200; and AND gates 202, 204 and 206. The combinational logicshown in FIG. 7 is merely exemplary, and persons skilled in the artunderstand that the signal NCTRL can be generated in various other ways.

With reference again to FIG. 5, in response to the signal PCTRL and afixed or constant PFET bias voltage, VB_P, a first stage PFET signalgenerator 208 produces a set of PFET control voltages, VB_P1, VB_P2,VB_P3, VB_P4, VB_P5, VB_P6 and VB_P7. Similarly, in response to thesignal NCTRL and a fixed or constant NFET bias voltage, VB_N, a firststage NFET signal generator 210 produces a set of NFET control voltages,VB_N1, VB_N2, VB_N3, VB_N4, VB_N5, VB_N6 and VB_N7.

As further illustrated in FIG. 8, first stage PFET signal generator 208includes six pairs of PFETs. In each pair, one PFET is controlled by oneof bits PCTRL2-PCTRL7 applied to the PFET gate, and the other PFET iscontrolled by the complement of that bit applied to the PFET gate.Similarly, as further illustrated in FIG. 9, first stage NFET signalgenerator 210 includes six pairs of NFETs. In each pair, one NFET iscontrolled by one of bits NCTRL2-NCTRL7 applied to the NFET gate, andthe other NFET is controlled by the complement of that bit applied tothe NFET gate.

As illustrated in FIG. 10, programmable-current first stage 134 receivesdifferential input signal VI (i.e., signals V1_N and V1_P), controlsignals VB_P and VB_N, a fixed or constant supply voltage VDD, a fixedor constant common-mode feedback voltage V_CMFB, and a fixed or constantcascode transistor bias voltage V_CASC and, in response, outputs thesignal V2 (i.e., signals V2_N and V2_P). Programmable-current firststage 134 includes a PFET branch 212, comprising 14 PFETs, and an NFETbranch 214 comprising 14 NFETs. The PFET branch 212 and NFET branch 214provide the controllable or programmable bias current to the amplifiertransistors 215. As noted above, the programmable-current first stage134 shown in FIGS. 5 and 10 is included in the in-phase (I) side of thesystem; an identical programmable-current first stage is included in thequadrature (Q) side of the system but is not shown for purposes ofclarity.

As illustrated in FIG. 11, programmable-current second stage 136receives differential input signal V2 (i.e., signals V2_N and V2_P),control signals AB0. AB1 and A2 and their complements, a fixed orconstant second-stage PFET bias voltage VAB_P, a fixed or constantsecond-stage NFET bias voltage VAB_N, and the supply voltage VDD. and,in response, outputs the signal V3 (i.e., signals V3_N and V3_P).Programmable-current second stage 136 includes three branches 216, 218and 220. Branch 216 remains in an “on” state (i.e., signal AB0 remainshigh) and contributes the same bias current in all instances ofoperation of the system, while branches 218 and 220 are used tocontribute the controllable or programmable portion of the bias current.Programmable-current second stage 136 shown in FIGS. 5 and 11 isincluded in the in-phase (I) side of the system; an identicalprogrammable-current second stage is included in the quadrature (Q) sideof the system but is not shown for purposes of clarity.

In operation, the output signal V3 represents the result of filteringthe input signal VI using TX-CTF system 74 (FIG. 4), where the biascurrent provided to the amplification circuitry of second amplifiersystem 120 is controlled or programmed in response to one or moretransmitter control signals.

The table in FIG. 12 illustrates an example of a relation between theISET signal, the resulting bias current provided to each of PFET branch212 and NFET branch 214 in programmable-current first stage 134 (FIG.10), and the total of the bias currents provided to these PFET and NFETbranches in the combined in-phase and quadrature (Q) sides of thesystem.

The table in FIG. 13 illustrates an example of a relation between the ABsignal, the resulting bias current provided to each of branches 218 and220 in programmable-current second stage 136 (FIG. 10), and the total ofthe bias currents provided to these branches in the combined in-phaseand quadrature (Q) sides of the system.

In the manner described above with regard to the exemplary embodiment,the present invention adjusts TX-CTF noise performance and linearity inresponse to transmitter operating conditions. To meet performancerequirements specified by the various standards (e.g., GMSK, EDGE,WCDMA, etc.) implicated by a multi-mode transceiver, it is desirable toprovide low noise and high linearity, but absent the present invention,simultaneously meeting all of the performance requirements of all of thevarious standards under all of the various operating conditions couldcome at the cost of high current consumption. To meet such performancerequirements with the lowest average power consumption, the presentinvention provides low noise and high linearity to no greater an extentthan demanded by transmitter operating conditions.

As described above, changing the bias current provided toprogrammable-current first stage 134 changes the noise in theamplification circuitry of programmable-current first stage 134. Higherbias current provided to programmable-current first stage 134 results inlow noise in the amplification circuitry of programmable-current firststage 134. Lower bias current provided to programmable-current firststage 134 results in higher noise in the amplification circuitry ofprogrammable-current first stage 134. The noise performance of theamplification circuitry of programmable-current first stage 134 is adominant contributor to the noise of TX-CTF system 74 as whole andultimately the noise of transmitter portion 68 (FIG. 2) as a whole. In,for example, WCDMA mode, the noise requirement is dependent ontransmitter output power. At higher transmit powers lower noise isrequired and vice versa. Also, in a back-off condition the noiserequirement is relaxed. By programming the programmable-current firststage 134 in response to one or more of modulation mode, transmit power,and back-off condition, TX-CTF system 74 provides the low noiseperformance only when required by such operating conditions, therebyminimizing the average power consumption of TX-CTF system 74 as a wholeand ultimately transmitter portion 68 as a whole.

Similarly, as described above, changing the bias current provided toprogrammable-current second stage 136 changes the output current driveand linearity in the amplification circuitry of programmable-currentsecond stage 136. Higher bias current provided to programmable-currentsecond stage 136 results in higher output current drive capability andhigher linearity in the amplification circuitry of programmable-currentsecond stage 136. Lower bias current provided to programmable-currentsecond stage 136 results in lower output current drive capability andlower linearity in the amplification circuitry of programmable-currentsecond stage 136. High current drive capability is required when highlinearity performance of TX-CTF system 74 is required, such as whentransmitter portion 68 is in GMSK mode, or when the input impedance ofdual-mode modulator and upconversion mixer 78 is lower, such as whentransmitter portion 68 is operating in a high frequency band.Conversely, a lower current drive capability is sufficient when lowerlinearity performance of TX-CTF system 74 is sufficient, such as whentransmitter portion 68 is in WCDMA mode, or when the input impedance ofdual-mode modulator and upconversion mixer 78 is higher, such as whentransmitter portion 68 is operating in a low frequency band. Byprogramming programmable-current second stage 136 in response to one orboth of transmitter mode and frequency band, TX-CTF system 74 providesthe high linearity performance only when required by such operatingconditions, thereby minimizing the average power consumption of TX-CTFsystem 74 as a whole and ultimately transmitter portion 68 as a whole.

While various embodiments of the invention have been described, it willbe apparent to those of ordinary skill in the art that many moreembodiments and implementations are possible that are within the scopeof this invention. For example, although examples of suitabletransmitter control signals and their use in determining an amount ofbias current to provide are described, others will occur readily topersons skilled in the art in view of the teachings herein. Accordingly,the invention is not to be restricted except in light of the followingclaims.

What is claimed is:
 1. A transmit continuous time filter system forattenuating aliasing and noise, comprising: a first programmable-currentcontinuous time filter stage having first current circuitry; and controllogic configured to provide first digital programming data to the firstprogrammable-current continuous time filter stage, the first currentcircuitry of the first programmable-current continuous time filter stageconfigured to generate a first level of programmable bias current inresponse the first digital programming data having a first value and togenerate a second level of programmable bias current lower than thefirst level in response to the first digital programming data having asecond value different from the first value.
 2. The system of claim 1wherein the first programmable-current continuous time filter stage ispart of a real-pole section of the transmit continuous time filtersystem.
 3. The system of claim 2 further including a biquadratic sectionhaving an output electrically connected to an input of the real-polesection.
 4. The system of claim 1 further comprising a secondprogrammable-current continuous time filter stage having second currentcircuitry, the control logic configured to provide second digitalprogramming data to the second programmable-current continuous timefilter stage.
 5. The system of claim 4 wherein the second currentcircuitry of the second programmable-current continuous time filterstage is configured to generate a third level of programmable biascurrent in response the second digital programming data having a thirdvalue and to generate a fourth level of programmable bias currentdifferent from the third level in response to the second digitalprogramming data having a fourth value different from the third value.6. The system of claim 1 wherein the control logic is configured togenerate the first digital programming data in response to one or moretransmitter control signals.
 7. The system of claim 6 wherein thecontrol logic is configured to set the first digital programming data tothe first value when the one or more transmitter control signalsindicate a first transmitter mode associated with a first noisethreshold and to set the first digital programming data to the secondvalue when the one or more transmitter control signals indicate a secondtransmitter mode associated with a second noise threshold lower than thefirst noise threshold.
 8. The system of claim 6 wherein the one or moretransmitter control signals include a transmitter power signalindicating one of at least a low transmit power or a high transmitpower, the control logic configured to set the first digital programmingdata to the first value when the transmitter power signal indicates thelow transmit power and to set the first digital programming data to thesecond value when the transmit power signal indicates the high transmitpower.
 9. The system of claim 6 wherein the one or more transmittercontrol signals include a transmitter band signal indicating one of atleast a low transmit frequency or a high transmit frequency, the controllogic configured to set the first digital programming data to the firstvalue when the transmitter band signal indicates the low transmitfrequency and to set the first digital programming data to the secondvalue when the transmitter band signal indicates the high transmitfrequency.
 10. The system of claim 6 wherein the one or more transmittercontrol signals include a transmitter modulation mode signal indicatingone of at least a wideband code-division multiple access mode or aGaussian Minimum Shift Keying mode, the control logic configured to setthe first digital programming data to the first value when thetransmitter modulation mode signal indicates the wideband code-divisionmultiple access mode and to set the first digital programming data tothe second value when the transmitter modulation mode signal indicatesthe Gaussian Minimum Shift Keying mode.
 11. The system of claim 6wherein the one or more transmitter control signals include a transmitpower back-off signal indicative of a transition from a high transmitpower mode to a low transmit power mode, the control logic configured toset the first digital programming data to the first value when thetransmit power back-off signal indicates the transition from the hightransmit power mode to the low transmit power mode.
 12. A radiofrequency subsystem for transmitting and receiving radio frequencysignals, comprising: a receiver portion configured to receive radiofrequency signals via an antenna; and a transmitter portion configuredto transmit radio frequency signals via the antenna, the transmitterportion including the transmit continuous time filter system of claim 1.13. The radio frequency subsystem of claim 12 further comprising afront-end module configured to allow passage of the radio frequencysignals from the transmitter portion to the antenna and from the antennato the receiver portion.
 14. The radio frequency subsystem of claim 12wherein the transmitter portion further includes a digital-to-analogconverter having an output electrically connected to an input of thetransmit continuous time filter system.
 15. The radio frequencysubsystem of claim 12 wherein the transmitter portion further includesan upconverter configured to upconvert a baseband signal from a basebandfrequency to a desired frequency for transmission, the upconverterhaving an input electrically connected to an output of the transmitcontinuous time filter system.
 16. The radio frequency subsystem ofclaim 12 wherein the transmitter portion further includes a poweramplifier system having a power amplifier and control circuitry, thecontrol circuitry configured to provide a signal to the transmitcontinuous time filter system, the signal indicative of a measured powerlevel at which the transmitter portion is operating.
 17. The radiofrequency subsystem of claim 12 wherein the control logic is configuredto provide first digital programming data in response to one or moretransmitter control signals indicating one of a first transmitter modeassociated with a first noise threshold and a second transmitter modeassociated with a second noise threshold lower than the first noisethreshold.
 18. A wireless communication device, comprising: an antenna;a baseband subsystem configured to output a digital baseband signal andone or more transmitter control signals; and a radio frequency subsystemconfigured to transmit an radio frequency signal based on the digitalbaseband signal via the antenna, the radio frequency subsystem includingthe transmit continuous time filter system of claim
 1. 19. The wirelesscommunication device of claim 18 wherein the one or more transmittercontrol signals indicate one of a first transmitter mode associated witha first noise threshold and a second transmitter mode associated with asecond noise threshold lower than the first noise threshold.
 20. Thewireless communication device of claim 18 wherein the baseband subsystemis configured to provide one or more of (i) a transmitter power signalindicating at least a low transmit power or a high transmit power, (ii)a transmitter band signal indicating at least a low transmit frequencyor a high transmit frequency, (iii) a transmitter modulation mode signalindicating at least a wideband code-division multiple access mode or aGaussian Minimum Shift Keying mode, and (iv) a transmit power back-offsignal indicative of a back-off condition in which the transmitterportion transitions from a high transmit power mode to a low transmitpower mode.